Compact dual-channel transceivers

ABSTRACT

In one embodiment, a method for monitoring a structure includes associating a passive sensor with the structure, the sensor being configured to measure a physical parameter of the structure that is indicative of the structure&#39;s condition, associating a dual-channel transceiver with the structure, the transceiver having a sensing repeater that operates on a first channel and a reference repeater that operates on a second channel, wherein the sensing repeater is connected to the passive sensor, receiving with a receive antenna of the reference repeater an interrogation signal transmitted from an interrogation device, multiplying a frequency of the received interrogation signal with the reference repeater to generate a reference response signal having a multiplied frequency, and transmitting the reference response signal with the reference repeater to the interrogation device.

CROSS-REFERENCE TO RELATED APPLICATION

This application is a divisional of co-pending U.S. Non-Provisionalapplication entitled, “Compact Dual-Channel Transceivers,” having Ser.No. 14/090,277, filed Nov. 26, 2013, which is hereby incorporated byreference herein in its entirety.

NOTICE OF GOVERNMENT-SPONSORED RESEARCH

This invention was made with Government support under grant contractnumbers 0925929 and 1232183 awarded by the National Science Foundation(NSF). The Government has certain rights in the invention.

BACKGROUND

Our society is increasingly dependent upon larger and more complexnetworks of civil infrastructure that are costly to maintain. Theseinfrastructures, especially those for transportation, are deterioratingdue to factors such as corrosion of the steel reinforcement and highcontinuous vibration. This type of degradation has motivated researchactivity on distributed wireless sensors that enable low-cost, nearcontinuous, and near real-time, non-destructive monitoring.

In order to efficiently provide such monitoring capabilities, it ispreferred to have the sensor node embedded within the structure indirect proximity of the physical or chemical activity of interest.Embedding the sensor node requires the device to be passive or rechargedexternally to extend the lifetime because of the difficultaccessibility, and to have a compact geometry to avoid compromising thestructural integrity and to facilitate the deployment of the node intothe preferred location. In order to communicate with embedded sensornodes over a long distance and from a convenient location, the sensingdevice should require low radio frequency (RF) activation power and beenergy efficient.

Another challenge associated with passive embedded wireless monitoringis remote channel calibration. When the sensor node is deployed in astructure, it is of concern to periodically calibrate the propagationchannel (between the interrogator and the node) to obtain a properreading from the sensor and confirm that the node is functioningproperly. Although such calibration can be obtained by placing areference node in close proximity to the sensor node, the requirementfor an additional node to be placed close to each sensing node isundesirable.

From the above discussion, it can be appreciated that it would bedesirable to have alternative apparatus that enables remote channelcalibration of an embedded sensor node.

BRIEF DESCRIPTION OF THE DRAWINGS

The present disclosure may be better understood with reference to thefollowing figures. Matching reference numerals designate correspondingparts throughout the figures, which are not necessarily drawn to scale.

FIG. 1 is a perspective view of an embodiment of a compact dual-channeltransceiver for use in embedded sensing applications.

FIG. 2A is a perspective view of a first (sensing) repeater of thetransceiver of FIG. 1.

FIG. 2B is a plan view of the first repeater of FIG. 2A.

FIGS. 3A and 3B are graphs that show (A) the simulated reflectioncoefficient between a receive antenna and a multiplier input and (B)between a transmit antenna and a multiplier output of the sensingrepeater at −30 dBm input power and 0 V bias.

FIGS. 4A and 4B are the simulated E- and H-plane radiation patterns of(A) the receive antenna at 2.4 GHz and (B) the transmit antenna at 4.8GHz.

FIG. 5A is a perspective view of a second (reference) repeater of thetransceiver of FIG. 1.

FIG. 5B is a plan view of the second repeater of FIG. 5A.

FIGS. 6A and 6B are graphs that show (A) the measured conversion gain(CG) of a dual-channel transceiver versus frequency for differentreceived powers and (B) the simulated and measured CG versus receivedpower at an f₁ of 2.4 GHz and 2.75 GHz.

FIG. 7A is a perspective view of a dual-channel transceiver similar tothat of FIG. 1, shown connected to a sensor.

FIG. 7B is a graph that shows the measured CG versus time for modulationsignals of frequency of 10 Hz and different amplitudes at an f₁ of 2.4GHz and received power of −30 dBm for the transceiver shown in FIG. 7A.

FIG. 8A illustrates a calibration measurement setup that was used forevaluation of a dual-channel transceiver.

FIG. 8B is a graph that shows the measured return power for differentvalues of f₁ versus bias voltage applied to the sensing repeater at −30dBm input power for the setup of FIG. 8A.

FIGS. 9A and 9B are graphs that show (A) the variation in extractedsensor voltage versus ΔP_(r) for different received power due to angleof incidence uncertainty and (B) δP_(r) for different azimuth angle andreceived power at 0 V bias.

FIGS. 10A and 10B illustrate an embedded sensor experimental setup.

DETAILED DESCRIPTION

As described above, it would be desirable to have apparatus that enablesremote channel calibration of an embedded sensor node. Disclosed hereinis a sensor node that has integrated means for calibrating the node. Insome embodiments, the sensor node comprises a compact three-dimensionaldual-channel transceiver that includes separate harmonic repeaters forsensing and providing a reference signal so as to enable remotecalibration and identification. In some embodiments, each repeatercomprises separate receive and transmit antennas that are coupled to afrequency multiplier that multiplies the frequency of an interrogationsignal that is received by the receive antenna. The sensing repeater iscoupled to a sensor that modulates the interrogation signal so that aresponse signal is reflected back to the interrogation device that isindicative of a sensed parameter (e.g., temperature, vibration, etc.).

In the following disclosure, various specific embodiments are described.It is to be understood that those embodiments are exampleimplementations of the disclosed inventions and that alternativeembodiments are possible. All such embodiments are intended to fallwithin the scope of this disclosure.

FIG. 1 illustrates an embodiment of a compact three-dimensionaldual-channel transceiver 10. In some embodiments, the transceiver 10 isconfigured to fit within the volume of a sphere 12 having a diameter ofapproximately 1.5 to 4 cm (e.g., 3 cm). As shown in FIG. 1, thetransceiver 10 generally comprises a first harmonic repeater 14, whichcan be used for sensing (i.e., a sensing repeater), and second harmonicrepeater 16, which can be used to provide a reference signal (i.e., areference repeater). The first repeater 14 comprises a base substrate 18as well as first and second antenna supports 20 and 22 that extendupward (vertically) from the base substrate. In some embodiments, thesupports 20, 22 are substrates having the same construction as the basesubstrate 18 and lie in planes that are orthogonal to the plane in whichthe base substrate lies. By way of example, the substrates can beconstructed of Rogers/RT Duroid 6006 material and have a nominalrelative dielectric constant (Er) of 6.5 and a thickness of 50 mils(1.27 mm). As described below, each support 20, 22 supports an antennaof the repeater 14.

As is further indicated in FIG. 1, formed on the base substrate 18 arefirst and second feeding networks 24 and 26 that are coupled to theantennas formed on the antenna supports 20, 22, respectively. Joiningthe feeding networks 24, 26 is a frequency multiplier 28, thatmultiplies the frequency of an input signal, such as a receivedinterrogation signal. In some embodiments, the frequency multiplier 28comprises a GaAs Schottky diode (e.g., Agilent HSCH-9161) that doublesthe frequency of the input signal. In such cases, the frequencymultiplier 28 can be referred to as a frequency doubler. As is alsoshown in FIG. 1, also formed on the base substrate 18 is a directcurrent (DC) bias network 30 that is coupled to the second feedingnetwork 26.

The second repeater 16 is similar in configuration to the first repeater14. Accordingly, the second repeater 16 generally comprises a basesubstrate 32 as well as first and second antenna supports 34 and 36 thatextend upward (vertically) from the base substrate (see FIG. 5A). Insome embodiments, the supports 34, 36 are substrates having the sameconstruction as the base substrate 32 and lie in planes that areorthogonal to the plane in which the base substrate lies. By way ofexample, the substrates can be constructed of Rogers/RT Duroid 6006material and have a nominal relative dielectric constant (Er) of 6.5 anda thickness of 50 mils (1.27 mm). Irrespective of their construction,each support 34, 36 supports an antenna of the repeater 16.

As is further shown in FIG. 1, the first repeater 14 and the secondrepeater 16 are physically connected together to form an integrateddevice. As indicated in the figure, the base substrates 18, 32 can beconnected to each other so that the first and second repeaters 14, 16have an inverted orientation relative to each other. In the orientationshown in FIG. 1, the antenna supports 20, 22 extend in an upwarddirection and the antenna supports 34, 36 extend in an opposite,downward direction.

FIGS. 2A and 2B show the first (sensing) repeater 14 in greater detail.More particularly, FIG. 2A shows the repeater 14 in a perspective viewand FIG. 2B shows the repeater in a top (plan) view. As is apparent fromFIG. 2A, each antenna support 20, 22 is generally triangular in shapeand therefore has a relatively wide base that narrows in the directionaway from the base substrate 18 because of angled edges of the support.In the illustrated embodiment, the first antenna support 20 is largerthan the second antenna support 22. By way of example, the first antennasupport 20 can be approximately 12 mm tall and its distal tip can beapproximately 8 mm wide, while the second antenna support 22 can beapproximately 10 mm tall and its distal tip can be approximately 7 mmwide.

Each antenna support 20, 22 comprises an inner side upon which anantenna is formed. Specifically, a receive antenna 40 is formed on theinner side of the first antenna support 20 and a transmit antenna 42 isformed on the inner side of the second antenna support 22. In theillustrated embodiment, each antenna comprises a meandered monopoleantenna including a meandered conductor trace 44 and 46 that is widernear the base (near the base substrate 18) and narrower near the tip(away from the base substrate).

FIG. 2B shows the feeding networks 24 and 26 of the first repeater 14more clearly. As indicated in this figure, the first feeding network 24comprises a bias network that includes a bypass capacitor 48 and an RFchoke 50. Extending downward through the base substrate 18 to a groundplane 52 formed on the bottom of the base substrate 18 (FIG. 2A) is avia 54. The second feeding network 26 also comprises a via 56 thatextends downward through the base substrate 18 to the ground plane 52.In addition, the second feeding network 26 comprises a shunt-shortedstub 64 that includes a via 56 that extends downward through the basesubstrate 18 to the ground plane 52.

In some embodiments, the first repeater 14 operates by receiving asignal at an f₁ of 2.4 GHz and transmitting a return signal at 2·f₁ of4.8 GHz. With further reference to FIG. 2B, the receive antenna 40 isfed by a microstrip line 60 that can have a characteristic impedance(Z_(o)) of 76Ω and electrical length of λ_(g)/16. The transmit antenna42 is fed by a microstrip line 62 that can have a length of λ_(g)/13 andZ_(o) of 74Ω and a λ_(g)/20 shunt-shorted stub 64 that can have a Z_(o)of 63Ω. The bias network 30 is added to provide an interface with theembedded sensor (not shown), as is discussed below. By way of example,the bias network 30 can comprise a 27 nH series inductor (Coilcraft0402) and 8.2 pF shunt capacitor (Johanson 0201).

An Advanced Design System (ADS) 2009u1 schematic was used to simulatethe frequency multiplier 28 and find the optimum impedances that theantennas 40, 42 need to present to the multiplier input/output toprovide the best conversion gain (CG) at −30 dBm input power. The CG isdefined herein as the ratio of the return power at 4.8 GHz to thereceived power at 2.4 GHz. Based on the simulation, it was found thatthe receive antenna 40 needs to present an impedance of 93+j358Ω at 2.4GHz and the transmit antenna 42 needs to present an impedance of25+j200Ω at 4.8 GHz. To provide these impedances, the High FrequencyStructure Simulator (HFSS) 14 was used to optimize parameters such asthe number of the monopole meandered sections, width of the sections,spacing of the sections, and the feeding network parameters. Thecomplete transceiver configuration shown in FIG. 1 was used for thesesimulations. A λ_(g)/10 64Ω shunt-shorted stub 30 was added to themultiplier circuit to fine tune the impedance match. This short-shortedstub 30 also provides a DC path for the diode return current. FIGS. 3Aand 3B show the reflection coefficient between the receive antenna 40and the multiplier input, and the transmit antenna 42 and the multiplieroutput, respectively, at −30 dBm input power. As shown in those figures,the reflected power at the fundamental frequency is very low.

One of the antenna design goals was to minimize size withoutsignificantly degrading the H-plane gain variation (the omni-directionalpattern) to facilitate interrogation. This was achieved by minimizingthe size of the antenna supports 20, 22 supporting the antennas 40, 42to reduce edge diffraction and by reducing the coupling between theantennas. The coupling was reduced by optimizing (1) the size of theground plane 52, (2) the distance between the antennas 40, 42, and (3)the height of the antennas. In addition, the meander pattern of theantenna traces 44, 46 and the location of each antenna 40, 42significantly contribute to the coupling. As shown in FIG. 1, thereceive and transmit antennas 40, 42 are positioned at opposite ends ofthe base substrate 18 and the meandering directions were chosen so thatthe induced near-field currents on each parasitic antenna oppose thecurrent direction on each driven antenna, thus self-canceling unwantedradiated far-fields. Reducing the coupling provides the additionalbenefit of having the reference signal independent of the modulatedreturn signal.

FIGS. 4A and 4B show the simulated radiation patterns of the receive andtransmit antennas 40, 42 at 2.4 GHz and 4.8 GHz, respectively. Theantennas 40, 42 are linearly polarized along the Z-axis relative to thecoordinate system shown in FIG. 2A. As shown in FIGS. 4A and 4B, theantennas 40, 42 demonstrate omni-directional patterns in the XY plane.The gain variation over the receive antenna H-plane is 1.6 dB, while itis 2 dB for the transmit antenna 42. The receive antenna peak gain is0.9 dB with a simulated radiation efficiency of 78.5%. The peak gain andradiation efficiency of the transmit antenna 42 are 1.1 dB and 67%,respectively. The distortion shown in the pattern, especially theE-plane, is due to (1) the relatively small ground plane size, (2)coupling from the other antennas, and (3) operating the antenna offresonance. For the same reasons, the radiation efficiency and gain weredegraded.

FIGS. 5A and 5B show the second (reference) repeater 16 in greaterdetail. More particularly, FIG. 5A shows the repeater 16 in aperspective view and FIG. 5B shows the repeater in a top (plan) view. Asis apparent from FIG. 5A, each antenna support 34, 36 is generallytriangular in shape and therefore has a relatively wide base thatnarrows in the direction away from the base substrate 32 because ofangled edges of the support.

In the illustrated embodiment, the first antenna support 34 is largerthan the second antenna support 36. By way of example, the first antennasupport 34 can be approximately 14 mm tall and its distal tip can beapproximately 8 mm wide, while the second antenna support 22 can beapproximately 11 mm tall and its distal tip can be approximately 7 mmwide.

Each antenna support 34, 36 comprises an inner side upon which anantenna is formed. Specifically, a receive antenna 70 is formed on theinner side of the first antenna support 34 and a transmit antenna 72 isformed on the inner side of the second antenna support 36. In theillustrated embodiment, each antenna 70, 72 comprises a meanderedmonopole antenna including a meandered conductor trace 74 and 76 that iswider near the base (near the base substrate 32) and narrower near thetip (away from the base substrate).

FIG. 5B shows feeding networks 78 and 80 that are coupled to the firstand second antennas 70, 72, respectively. As indicated in this figure,the first feeding network 78 comprises a bias network that includes abypass capacitor 82 and an RF choke 84. Extending downward through thebase substrate 18 to a ground plane 86 formed on the bottom of the basesubstrate 32 (FIG. 5A) is a via 88. The second feeding network 80 alsocomprises a via 90 that extends downward through the base substrate 32to the ground plane 86. Joining the feeding networks 78, 80 is afrequency multiplier 92, that multiplies the frequency of an inputsignal. In some embodiments, the frequency multiplier 92 comprises aGaAs Schottky diode (e.g., Agilent HSCH-9161) that doubles the frequencyof the input signal. As is also shown in FIG. 5B, the second feedingnetwork 80 comprises a shunt-shorted stub 104 that includes a via 90that extends down to the ground plane 86.

In some embodiments, the second repeater 16 is designed to receive asignal at an f₁ of 2.75 GHz and transmit a signal back at 2·f₁ of 5.5GHz. Based on simulated data, the frequency separation between the first(sensing) repeater 14 and the second (reference) repeater 16 can bereduced to 200 MHz at the fundamental frequency without a significantincrease in the coupling between the antennas. If smaller frequencyspacing is required, the distance between the antennas and the groundplane size can be increased. The receive antenna 70 is fed by a λ_(g)/1469Ω microstrip line 98 and a 3.9 nH series inductor 100 (e.g., Coilcraft0402). The transmit antenna 76 is fed by a λ_(g)/11 69Ω microstrip line102 and a λ_(g)/18 shunt-shorted stub 104 with a Z_(o) of 64Ω. Similarto the multiplier circuit of the first repeater 14, an additionalshunt-shorted stub 94 of an electrical length of λ_(g)/9 and Zo of 64Ωwas added to fine tune the impedance match. In the case of the secondrepeater 16, the bias network is included in the design for measurementpurposes only.

Based on the ADS simulation, it was found that the receive and transmitantennas 70, 72 should present an impedance of 67+j250Ω at 2.75 GHz andan impedance of 30+j160Ω at 5.5 GHz, respectively. The referencerepeater reflection coefficients between the antennas 70, 72 andmultiplier circuit, as well as the radiation patterns, are similar tothose of the first repeater 14. The variation over the receive antennaH-plane is 1 dB, while it is 0.8 dB for the transmit antenna 72. Thereceive antenna peak gain is 1.8 dB with a simulated radiationefficiency of 94.5%. The peak gain and radiation efficiency of thetransmit antenna 72 are 2 dB and 98%, respectively. The antennas 70, 72have better gain and efficiency performance than those of the firstrepeater 14 because of the relative increase in the occupied electricalsize.

The performance of the transceiver design described above wascharacterized inside an anechoic chamber. The measurements wereperformed by transmitting a signal at an f₁, which was generated by avector network analyzer (VNA), and measuring the return signal at 2·f₁using a spectrum analyzer. The transmit and receive interrogatorantennas were placed at a distance of 1 m from the transceiver undertest and separated by 1.5 m. Low-pass filters were used in the transmitside to filter out the VNA-generated harmonics. FIGS. 6A and 6B show themeasured and simulated CG performance for different input power levelsand f₁. The simulated CG of the entire repeater was calculated by addingthe sum of antenna gains to the simulated CG curve of the multiplieralone. For an input power of −30 dBm, the CG maxima are −15.5 dB at anf₁ of 2.4 GHz, and −15.7 dB at an f₁ of 2.75 GHz. The 3 dB CG bandwidthis 2% and 1.25% at f₁ of 2.4 GHz and input powers of −20 dBm and −30dBm, respectively. At an f₁ of 2.75 GHz, the 3 dB CG bandwidth is −2.5%for input powers of −20 dBm and −30 dBm. As shown in FIG. 6B, themeasured CG versus received power at an f₁ of 2.4 GHz is well matchedwith the simulated curve while there is a difference between themeasured and simulated curves at an f₁ of 2.75 GHz at low power. Thisdifference can be attributed to fabrication and assembly errors.

Integration of a sensor into the transceiver node was also studied. Inorder to illustrate the ability to modulate the return signal, afunction generator was used to apply a sinusoidal 10 Hz signal at thefrequency doubler input through the bias network and the power level ofthe return signal was measured versus time (FIG. 7B). This applied ACwaveform represents the signal that would be provided by a sensor. Thismeasurement was performed for an RF input power level of −30 dBm. Asshown in FIG. 7B, when the amplitude of the AC signal is set to 0 V, thereturn signal level is constant. By applying a low-level signal (20 mV),an amplitude modulated return signal can be detected. Increasing theamplitude of the AC signal increases the variation of the return signallevel. In order to determine the sensitivity of the transceiver tohousing a sensor within the structure, full-wave simulations wereperformed that included a physical model for a commercial piezo thinfilm vibration sensor (MiniSense 100NM from Measurement Specialties) asshown in FIG. 7A. The presence of the sensor causes a frequency shift of10 MHz and 1% reduction in the radiation efficiency of the receiveantenna and 1 dB increase in the gain variation over the transmitantenna H-plane. This degradation can be mitigated by re-optimizing thedesign while the sensor is included.

Table I shows a comparison of the maximum CG, operating power, andelectrical size (ka) of the presented transceiver design and othersingle-channel harmonic repeater tags from the prior art. Ka representsthe size of the overall structure, where k=2π/λ, λ=free space wavelengthat f₁ and a=radius of the smallest sphere enclosing the maximumdimension of the tag. As shown in Table I, although the proposed designcontains integrated two repeaters, the CG performance and overallelectrical occupied volume compare well with the compact, most efficientsingle channel designs. In addition, the proposed dual-channeltransceiver provides omnidirectional interrogation capability whichfacilitates the deployment of the node and the interrogation. Theexpected communication at f₁ of 2.4 GHz range using an interrogator withan effective isotropic radiated power (EIRP) of 2 W and 15 dBi gainantenna on the receive side is 45 m in free space.

TABLE I Comparison Of The CG And Electrical Size With Others FromLiterature RF Power f₁ Parameter Max CG (dB) (dBm) ka (GHz) PresentedDesign −7.2 −18 0.78 2.4 [13] −8.4 −20 0.75 2.4  [9] −19.7 −15 0.78 1.3[10] −35.4 −19.5 076 1.3 [11] −11 −41 0.83 5.9

Calibration of the propagation channel becomes a practical challenge forthe implementation of a long-range passive embedded monitoring system.Such calibration is required when the exact characteristics of theinterrogator-node link are unknown or vary with time. When the signalthat is returned from the sensing repeater slowly changes over time, aswould occur for a temperature sensor, the reference repeater can be usedto provide a constant signal level to compare with the sensor activity.For the sensors that provide a relatively high-frequency modulatedreturn signal, the reference signal can be used to ensure that the nodeis operable and to determine the location of the node. For both of thesescenarios, the reference signal can also be used to identify the node.

FIG. 8B illustrates the measured return power level at an f₁ of 2.4 GHzand an f₁ of 2.75 GHz for different applied bias voltage to the sensingrepeater and different received power. This measurement was performed asillustrated in FIG. 8A. As shown in FIG. 8B, when interrogating thetransceiver with a signal frequency of 2.75 GHz, the reference repeaterprovides a nearly constant return signal at 2·f₁ of 5.5 GHz that isunaffected by the bias on the sensing repeater. The variation of the 5.5GHz signal amplitude is only +/−0.5 dB and +/−0.3 dB for RF input powerof −20 and −30 dBm, respectively. Conversely, interrogating thetransceiver with a signal frequency f₁ of 2.4 GHz provides a returnsignal at 2·f₁ of 4.8 GHz that is highly sensitive to the applied DCvoltage; 0.1 V bias decreases the return signal by >16 dB. Bysubtracting the amplitude of the 5.5 GHz return signal from the 4.8 GHzreturn signal, the modulated return signal can be corrected for channeleffects, in practice providing a calibrated sensor reading. Thisdifference between the return signal from the reference and sensingrepeaters is referred to as ΔP_(r).

Because the H-plane radiation patterns at the sensing and referencerepeater frequencies have a gain difference that changes versus angle,the calibrated sensor reading can have uncertainty. That is, in additionto the sensor activity, ΔP_(r) will vary depending on the angle ofincidence (φ) of the interrogation signal, which may not be known tohigh accuracy. Based on the simulated radiation patterns, the maximumAP, variation (δP_(r)) at 0 V bias versus angle is found to be −2.2 dBfor RF input power of −20 dBm and −30 dBm (FIG. 9B). Using the data inFIG. 8B, ΔP_(r) can be superimposed on ΔP_(r) to determine thecorresponding variation that would be found in the extracted value ofthe sensor-provided voltage (FIG. 9A). For sensors that generate voltagebetween 60-200 mV, the maximum voltage variation is 5 mV and 6.5 mV forRF input power of −30 dBm and −20 dBm, respectively. This variationassumes that the angle of incidence may vary between 0° and 360°,however, in an actual application the angular uncertainty may be muchless than 45°. For simplification, it was assumed in this assessmentthat the sensor provides only positive voltages.

A method for identifying individual nodes, when no or very limitedsignal processing is possible at the node, was also investigated.Different approaches can be used to address this challenge, such asintroducing time or phase delay between the nodes or through frequencydiversity. Due to potentially severe multi-path propagationenvironments, however, the frequency diversity method is more viable forembedded sensing. With the disclosed dual-channel transceiver,identification can be performed if a unique design frequency is chosenfor the reference repeater of each node. In operation, the interrogationfrequency can be swept until the return signal peaks and the node isidentified. Given the narrowband characteristics of the nodes, theidentification signal from the interrogator is easily isolated from thesensor activity. Furthermore, both the reference and sensing repeaterfrequencies can be selectable, which squares the number of unique nodeidentifications (IDs). The use of high Q filters to further reduce theCG bandwidth of the repeater would noticeably degrade the CG efficiency.

One approach that can further increase the ID count without affectingperformance is to combine polarization and frequency diversity. Theantennas used in the transceiver are linearly polarized and, whendeployed, adjacent nodes can be oriented to have a given amount ofpolarization mismatch. A conservative estimate is that 5 dB of mismatchwould be sufficient to distinguish between adjacent nodes. Due to thenon-linear CG versus power slope, the measured co-pol to cross-pol ratiowas found to be at least 40 dB. Therefore, polarization diversity canincrease the number of IDs by a factor of 8.

Table II provides the expected number of IDs assuming the interrogatoroperating bandwidth is 2-3 GHz and the amplitude isolation between thenodes is required to be 5 dB. As shown in the table, by changing thenode polarization and the reference repeater frequency, the number ofnodes is 48 and 56 for RF power levels of −20 dBm and −30 dBm receivedat the node, respectively. These numbers are calculated using theinterrogator bandwidth divided by the repeater bandwidth (both in MHz)multiplied by 8 for the polarization diversity factor. Cases with <200MHz separation between the sensing and reference frequencies wereexcluded. Changing both the reference and sensing repeater operatingfrequencies and the polarization will increase the number of IDs to 768and 2112 for RF power levels of −20 dBm and −30 dBm, respectively.Employing frequency diversity to both channels has the additionalsignificant benefit of allowing different nodes located within theinterrogator beam angle to be identified, thus overcoming thelimitations on the distance between the deployed nodes, interrogationdistance, and interrogator antenna design (directivity, beam width andsize).

TABLE II Number Of Node IDs Using A 2-3 GHz Interrogator For DifferentRF Power Levels (With Polarization Diversity) −20 dBm −30 dBm 5 dB CG BW(sensing repeater) 5% 3.2% 5 dB CG BW (reference repeater) 3% 1.6% Minno. of nodes, changing reference  48  56 repeater frequency Min no. ofnodes, changing sensing 768 2112 and reference repeater frequency

A preliminary study of the impact of embedding the proposed transceiverwas also performed. FIGS. 10A and 10B illustrate the experimental setupthat was used. The transceiver is buried in the middle of a box whichwas filled with dry sand and has four cylindrical metal bars. The metalbars were inserted to better represent a structure with rebarenforcement. The side dimension of the sand box was 30 cm and the barshad a diameter of 1 cm and height of 20 cm. To protect the transceiverfrom damage and to reduce the effect of the sand direct dielectricloading, the transceiver was placed inside a 5×5×5 cm³ foam box(electrical properties similar to air). During the measurement, thetransceiver was oriented to achieve the maximum CG. The CG maximaoccurred at shifted f₁ frequencies of 2.39 GHz and 2.72 GHz with similarbandwidths to what was measured in air. The frequency shifts can becorrected for by optimizing the design to operate in a sand environment.The embedding also caused degradation in the CG by 1 dB (at 2.39 GHz)and 6 dB (at 2.72 GHz) for −20 dBm input power. The degradation wasapproximately 2 dB higher at each frequency for −30 dBm input power.This degradation was due to attenuation through the sand layer and thefrequency shift that cause differences in the impedance matchconditions. For these reasons the degradation at 2.72 GHz is muchhigher. In addition, the embedding experiment illustrated a decrease inthe co-pol to cross-pol isolation by 8 dB. This decrease was due tosignal diffraction at the air/sand interfaces and multi-path reflectionsof the experiment setup.

The effect of the surrounding medium on the frequency of the CG maximaand the polarization of the return signal will cause uncertainty in thenode ID and will require the frequency spacing and polarization mismatchlevel between the IDs to be larger. For certain sensing environments inwhich the multi-path and interference levels are high, difficulties inthe proposed remote channel calibration approach will also beintroduced.

The invention claimed is:
 1. A method for monitoring a structure, themethod comprising: associating a passive sensor with the structure, thesensor being configured to measure a physical parameter of the structurethat is indicative of the structure's condition; associating adual-channel transceiver with the structure, the transceiver having asensing repeater that operates on a first channel and a referencerepeater that operates on a second channel, wherein the sensing repeateris connected to the passive sensor; receiving with a receive antenna ofthe reference repeater an interrogation signal transmitted from aninterrogation device; multiplying a frequency of the receivedinterrogation signal with the reference repeater to generate a referenceresponse signal having a multiplied frequency; and transmitting thereference response signal with the reference repeater to theinterrogation device.
 2. The method of claim 1, wherein associating asensor with a structure comprises associating with the structure asensor configured to measure temperature or vibration of the structure.3. The method of claim 1, wherein associating a dual-channel transceiverwith the structure comprises embedding the dual-channel transceiverwithin the structure.
 4. The method of claim 1, wherein multiplying afrequency of the received interrogation signal comprises multiplying thefrequency with a harmonic repeater.
 5. The method of claim 1, whereinmultiplying a frequency of the received interrogation signal comprisesdoubling the frequency.
 6. The method of claim 5, wherein doubling thefrequency comprises doubling the frequency with a Schottky diode.
 7. Themethod of claim 1, further comprising receiving the reference responsesignal with the interrogation device and calibrating a propagationchannel between the interrogation device and the dual-channeltransceiver with the response signal.
 8. The method of claim 1, furthercomprising receiving the reference response signal with theinterrogation device and determining a location of the dual-channeltransceiver using the response signal.
 9. The method of claim 1, furthercomprising: receiving with a receive antenna of the sensing repeater asecond interrogation signal transmitted from the interrogation device;multiplying a frequency of the received second interrogation signal onthe sensing repeater to generate a second response signal; modulatingthe second response signal with an output from the sensor; andtransmitting a modulated response signal with the sensing repeater tothe interrogation device.
 10. The method of claim 9, wherein multiplyinga frequency of the received second interrogation signal comprisesmultiplying the frequency with a harmonic repeater.
 11. The method ofclaim 9, wherein multiplying a frequency of the received secondinterrogation signal comprises doubling the frequency.
 12. The method ofclaim 11, wherein doubling the frequency comprises doubling thefrequency with a Schottky diode.
 13. The method of claim 9, furthercomprising determining a physical parameter of the structure from themodulated response signal.